Radar front-end with rf oscillator monitoring

ABSTRACT

A radar method is described. According to one exemplary embodiment, the method includes generating a first RF oscillator signal in a first chip and supplying the first RF oscillator signal to a transmission (TX) channel of the first chip and transmitting the first RF oscillator signal from the TX channel of the first chip to the second chip via a transmission line.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.16/105,111 filed on Aug. 20, 2018, which claims the benefit of GermanPatent Application No. 102017118987.4 filed on Aug. 18, 2017 and GermanPatent Application No. 102018117688.0 filed on Jul. 22, 2018, which areincorporated by reference as if fully set forth.

FIELD

The present description relates to the field of radio-frequency (RF)circuits. Some exemplary embodiments relate to a radar chip which issuitable for being cascaded with another radar chip.

BACKGROUND

Radio-frequency (RF) transmitters and receivers are found in amultiplicity of applications, in particular in the field of wirelesscommunication and radar sensors. In the automotive sector, there is anincreasing need for radar sensors which are used in so-called adaptivecruise control (ACC, or Radar Cruise Control) systems. Such systems canautomatically adapt the speed of an automobile in order to thus maintaina safe distance to other automobiles traveling in front (and from otherobjects and pedestrians). Further applications in the automotive sectorare, for example, blind spot detection, lane change assist and the like.

Modern radar systems use highly integrated RF circuits which can combineall core functions of an RF front-end of a radar transceiver in a singlehousing (single-chip radar transceiver), which is often referred to as amonolithic microwave integrated circuit (MMIC). Such RF front-endsusually comprise, inter alia, a voltage-controlled oscillator (VCO)connected in a phase-locked loop, power amplifiers (PA), directionalcouplers, mixers and analog-to-digital converters (ADC) as well asassociated control circuit arrangements for controlling and monitoringthe RF front-end.

Modern frequency-modulated continuous-wave (FMCW) radar systems areoften multi-input/multi-output (MIMO) systems having a plurality oftransmission (TX) and reception (RX) channels. MIMO systems usuallycomprise a plurality of MMICs which are arranged on a carrier board(PCB, printed circuit board) and must operate in a synchronous manner,wherein each MMIC per se may have a plurality of RX and TX channels. Oneobject may be considered to be the (in-phase) synchronization of an MIMOradar system having a plurality of MMICs.

SUMMARY

A radar method is described. According to one exemplary embodiment, themethod includes generating a first RF oscillator signal in a first chipand supplying the first RF oscillator signal to a transmission (TX)channel of the first chip and transmitting the first RF oscillatorsignal from the TX channel of the first chip to the second chip via atransmission line.

According to another exemplary embodiment, the method includesgenerating a first RF oscillator signal in a first chip and supplyingthe first RF oscillator signal to a transmission (TX) channel of thefirst chip. The method also includes generating a second RF oscillatorsignal in a second chip and transmitting the first RF oscillator signalfrom the first chip to the second chip via a transmission line. Themethod also includes determining a first propagation delay of the firstRF oscillator signal arriving at the second chip by means ofdemodulation using the second RF oscillator signal.

A radar chip is also described. According to one exemplary embodiment,the radar chip has: a local oscillator (LO) for generating an RFoscillator signal and at least one first TX channel which is connectedto a pin of the MMIC. The TX channel can be configured both to outputthe RF oscillator signal via the pin of the first TX channel and toreceive a further RF oscillator signal via the pin of the first TXchannel.

A radar system is also described. According to one exemplary embodiment,the radar system includes a first chip and at least one second chip,wherein the first chip has an RF oscillator which is designed togenerate an RF oscillator signal and to output it at a first RF outputcontact. The system also includes an RF splitter which is arranged onthe carrier and has an input and a first output and at least one secondoutput. A first transmission line connects the RF output contact of thefirst chip to the input of the RF splitter. A second transmission lineconnects the first output of the RF splitter to an RF input of the firstchip and a third transmission line connects the second output of the RFsplitter to an RF input of the second chip. In this case, the second andthird transmission lines are configured in such a manner that they causethe same propagation delay during operation when transmitting the RFoscillator signal.

According to another exemplary embodiment, a radar system includes afirst chip having a first RF contact and a second chip having a secondRF contact as well as a first RF oscillator which is integrated in thefirst chip and has an output which is coupled to the first RF contactvia at least one transmission (TX) channel. The system also includes asecond RF oscillator which is integrated in the second chip, atransmission line which connects the first RF contact on the first chipto the second RF contact on the second chip and at least one firstdemodulator which is arranged in the second chip and has an RF inputwhich is coupled to the second RF contact and a reference input which iscoupled to an output of the second RF oscillator. The first RFoscillator is designed to generate a first RF oscillator signal which istransmitted to the RF input of the first demodulator via the first RFcontact, the transmission line and the second RF contact. A control unitis designed to determine a first propagation delay of the first RFoscillator signal arriving at the second chip on the basis ofinformation obtained from the first demodulator.

A method is also described and, according to one exemplary embodiment,includes: receiving a first RF oscillator signal via an RF chip contactof a radar chip; generating a second RF oscillator signal by means of anRF oscillator in the radar chip, wherein the first RF oscillator signaland the second RF oscillator signal have the same frequency and anadjustable phase shift relative to one another; generating a sum signalby superimposing the first RF oscillator signal and the second RFoscillator signal; and generating a plurality of measured values, whichrepresent the power of the sum signal, for a plurality of phase shifts,wherein a phase shift is assigned to each of the measured values.Finally, a value representing the power of the first RF oscillatorsignal is determined on the basis of the plurality of measured values.

Another exemplary embodiment relates to an RF circuit having an RF chipcontact of a radar chip for receiving an external first RF oscillatorsignal and a local oscillator which is arranged in the radar chip and isdesigned to generate a second RF oscillator signal, wherein the first RFoscillator signal and the second RF oscillator signal have the samefrequency. The RF circuit also has a phase shifter which is designed toset a phase shift between the first and second oscillator signals, andan RF power detector which is arranged in the radar chip. The powerdetector has an input which is coupled to the RF chip contact and to anoutput of the local oscillator by means of a coupler, with the resultthat the first RF oscillator signal and the second RF oscillator signalare superimposed at the input of the RF power detector, with the resultthat the power detector measures the power of the superimposed signal.

BRIEF DESCRIPTION OF THE DRAWINGS

Exemplary embodiments are explained in more detail below using figures.The illustrations are not necessarily true to scale and the exemplaryembodiments are not only restricted to the aspects illustrated. Rather,importance is placed on illustrating the principles on which theexemplary embodiments are based. In the figures:

FIG. 1 is a sketch for illustrating the functional principle of an FMCWradar system for measuring distance and/or speed.

FIG. 2 comprises two timing diagrams for illustrating the frequencymodulation of the RF signal generated by the FMCW system.

FIG. 3 is a block diagram for illustrating the fundamental structure ofan FMCW radar system.

FIG. 4 is a block diagram for illustrating an example of an analog RFfront-end of the FMCW radar system from FIG. 3.

FIG. 5 illustrates one example of an MIMO radar system having aplurality of cascaded MMICs.

FIG. 6 illustrates one example of a radar system in which a master MMICis coupled to at least one slave MMIC, wherein the local oscillatorsignal generated by the master MMIC is both supplied to the slave MMICand fed back to the master MMIC.

FIG. 7 illustrates an exemplary implementation of a radar system havinga plurality of transmission channels which are arranged in a master MMICwith optional feedback of the oscillator signal.

FIG. 8 illustrates a further exemplary implementation of a radar systemhaving a plurality of transmission channels which are arranged in amaster or slave MMIC with optional feeding of the local oscillatorsignal, wherein a TX channel can be configured as an input for the localoscillator signal.

FIGS. 9A-9C illustrate a radar system having two coupled MMICs, whereinat least one transmission channel respectively has a circuit fordetecting the phase and (optionally) frequency of the local oscillatorsignal, where FIG. 9A illustrates a signal flow from the master MMIC tothe slave MMIC and FIG. 9B illustrates the reverse signal flow; and FIG.9C illustrates a simplified version of FIG. 9B.

FIG. 10 illustrates one example of a measurement of the propagationdelay of the LO signal from a master MMIC to a slave MMIC.

FIG. 11 illustrates the measurement of the propagation delay of the LOsignal from a slave MMIC to a master MMIC, corresponding to FIG. 10.

FIG. 12 illustrates another example of the measurement of thepropagation delay of the LO signal from a master MMIC to a slave MMIC(and vice versa).

FIG. 13 illustrates another example of an MMIC which has substantiallythe same structure as the MMICs in FIGS. 9A-9C, but with an additionalphase modulator in the TX channel.

FIG. 14 illustrates an exemplary illustration of a method for detectingthe propagation delay or phase shift of an LO signal during transmissionfrom a first (for example master) MMIC to a second (for example slave)MMIC.

FIG. 15 illustrates a part of a radar system having two coupled MMICs,wherein only those components which are used to measure the power of thesignal received in the slave MMIC are illustrated.

FIG. 16 is a graph for illustrating the measured values obtained duringthe power measurement.

FIG. 17 is a flowchart for illustrating one example of a method formeasuring the power of an LO signal received in a (for example slave)radar chip.

DETAILED DESCRIPTION

The following exemplary embodiments here are described in the context ofa radar receiver. However, the various exemplary embodiments describedhere are not restricted to radar applications and can also be used inother areas, for example in RF transceivers of RF communicationapparatuses. RF circuits from the wide variety of fields of applicationmay have voltage-controlled oscillators (VCOs) for generating RFsignals. Instead of VCOs, it is alternatively also possible to usedigitally controlled oscillators (DCOs). The concepts described here canbe readily easily applied to applications in which DCOs are used insteadof VCOs.

FIG. 1 illustrates the use of an FMCW radar system as a sensor formeasuring distances and speeds of objects which are usually referred toas radar targets. In the present example, the radar apparatus 10 hasseparate transmission (TX) and reception (RX) antennas 5 and 6 (bistaticor pseudo-monostatic radar configuration). However, it is noted that itis also possible to use a single antenna which is simultaneously used asa transmission antenna and a reception antenna (monostatic radarconfiguration). The transmission antenna 5 emits a continuous RF signals_(RF)(t) which is frequency-modulated by means of a sawtooth signal(periodic linear ramp signal), for example. The emitted signal s_(RF)(t)is scattered back at the radar target T and the back-scattered(reflected) signal y_(RF)(t) is received by the reception antenna 6.

FIG. 2 illustrates, by way of example, the mentioned frequencymodulation of the signal s_(RF)(t). As illustrated in FIG. 2, the signals_(RF)(t) is composed of a set of “chirps”, that is to say the signals_(RF)(t) comprises a sequence of sinusoidal signal profiles (waveforms)with a rising (up-chirp) or falling (down-chirp) frequency (see uppergraph in FIG. 2). In the present example, the instantaneous frequencyf(t) of a chirp linearly rises at a starting frequency f_(START)beginning inside a period T_(RAMP) to a stop frequency f_(STOP) (seelower graph in FIG. 2). Such chirps are also referred to as a linearfrequency ramp. FIG. 2 illustrates three identical linear frequencyramps. However, it is noted that the parameters f_(START), f_(STOP),T_(RAMP) and the pause between the individual frequency ramps can vary.The frequency variation also need not necessarily be linear. Dependingon the implementation, transmission signals with an exponential(exponential chirps) or hyperbolic (hyperbolic chirps) frequencyvariation can also be used.

FIG. 3 is a block diagram which illustrates, by way of example, apossible structure of a radar apparatus 1 (radar sensor). Similarstructures can also be found in RF transceivers, for example, which areused in other applications, for example wireless communication systems.Therefore, at least one transmission antenna 5 (TX antenna) and at leastone reception antenna 6 (RX antenna) are connected to an RF front-end 10which can comprise all those circuit components which are required forRF signal processing. These circuit components comprise, for example, alocal oscillator (LO), RF power amplifiers, low-noise amplifiers (LNA),directional couplers (for example rat-race couplers, circulators, etc.)and mixers for down-mixing the RF signals to the baseband or anintermediate frequency band (IF band). The RF front-end 10 can beintegrated—possibly together with further circuit components—in amonolithic microwave integrated circuit (MMIC). The illustrated exampleshows a bistatic (or pseudo-monostatic) radar system with separate RXand TX antennas. In the case of a monostatic radar system, a singleantenna (or an antenna array) would be used both to emit and to receivethe electromagnetic (radar) signals. In this case, a directional coupler(for example a circulator) can be used to separate the RF signals to beemitted into the radar channel from the RF signals (radar echoes)received from the radar channel.

In the case of a frequency-modulated continuous-wave radar system (FMCWradar system), the RF signals emitted via the TX antenna 5 may be, forexample, in the range of approximately 20 GHz to 81 GHz (for example 77GHz in some applications). As mentioned, the RF signal received by theRX antenna 6 comprises the radar echoes, that is to say those signalcomponents which are scattered back at the so-called radar targets. Thereceived RF signal y_(RF)(t) is down-mixed to the baseband, for example,and is processed further in the baseband by means of analog signalprocessing (see FIG. 3, analog baseband signal processing chain 20).Said analog signal processing comprises substantially filtering andpossibly amplification of the baseband signal. The baseband signal isfinally digitized (see FIG. 3, analog-to-digital converter 30) andprocessed further in the digital range. The digital signal processingchain can be at least partially implemented as software which isexecuted on a processor (see FIG. 3, DSP 40). The overall system isgenerally controlled by means of a system controller 50 which canlikewise be at least partially implemented as software which can beexecuted on a processor, for example a microcontroller. The RF front-end10 and the analog baseband signal processing chain 20 (optionally alsothe analog-to-digital converter 30) can be integrated together in asingle MMIC (that is to say an RF semiconductor chip). Alternatively,the individual components can also be distributed among a plurality ofintegrated circuits.

FIG. 4 illustrates an exemplary implementation of the RF front-end 10with a downstream baseband signal processing chain 20 which may be partof the radar sensor from FIG. 3. It is noted that FIG. 4 illustrates asimplified circuit diagram in order to show the fundamental structure ofthe RF front-end. Actual implementations which may depend greatly on thespecific application may naturally be more complex. The RF front-end 10comprises a local oscillator 101 (LO) which generates an RF signals_(LO)(t). As described above with reference to FIG. 3, the signals_(LO)(t) may be frequency-modulated and is also referred to as an LOsignal. In radar applications, the LO signal is usually in the SHF(Super High Frequency, centimeter wave) or in the EHF (Extremely HighFrequency, millimeter wave) band, for example in the range of 76 GHz to81 GHz in automotive applications.

The LO signal s_(LO)(t) is processed both in the transmission signalpath and in the reception signal path. The transmission signal s_(RF)(t)(cf. FIG. 2) which is emitted by the TX antenna 5 is generated byamplifying the LO signal s_(LO)(t), for example by means of the RF poweramplifier 102. The output of the amplifier 102 can be coupled to the TXantenna 5 (in the case of a bistatic or pseudo-monostatic radarconfiguration). The reception signal y_(RF)(t) which is provided by theRX antenna 6 is supplied to the RF port of the mixer 104. In the presentexample, the RF reception signal y_(RF)(t) (antenna signal) ispreamplified by means of the amplifier 103 (gain g) and the amplified RFreception signal g·y_(RF)(t) is supplied to the mixer 104. The amplifier103 may be an LNA, for example. The LO signal s_(LO)(t) is supplied tothe reference port of the mixer 104, with the result that the mixer 104down-mixes the (preamplified) RF reception signal y_(RF)(t) to thebaseband. The down-mixed baseband signal (mixer output signal) isdenoted y_(BB)(t). This baseband signal y_(BB)(t) is initially furtherprocessed in an analog manner, wherein the analog baseband signalprocessing chain 20 has substantially amplification (amplifier 22) andfiltering (for example bandpass filter 21) in order to suppressundesired sidebands and image frequencies. The resulting analog outputsignal which can be supplied to an analog-to-digital converter isdenoted y(t). Methods for the digital further processing of the outputsignal (digital radar signal) are known per se (for example therange-Doppler analysis) and are therefore not discussed any furtherhere.

In the present example, the mixer 104 mixes the preamplified RFreception signal g·y_(RF)(t) (that is to say the amplified antennasignal) down to the baseband. The mixing can be carried out in one stage(that is to say from the RF band directly to the baseband) or via one ormore intermediate stages (that is to say from the RF band to anintermediate frequency band and then to the baseband). In view of theexample shown in FIG. 4, it becomes clear that the quality of a radarmeasurement depends greatly on the quality or accuracy and on the phaseangle of the LO signal s_(LO)(t). In particular, the phase of the LOsignal s_(LO)(t) supplied to the mixer 104 (as reference signal) is alsoimportant for an accurate measurement. In the case of multichannel radarsystems having a plurality of reception channels (RX channels), thephase angle of the oscillator signals supplied to the RX channels asreference signals has a significant influence on the measurement of theangle of incidence (Direction of Arrival, DoA) of the received radarsignals.

FIG. 5 is a block diagram which illustrates, by way of example, an MIMOradar system having a plurality of coupled (cascaded) MMICs. In theexample illustrated, three MMICs are arranged on a carrier, for examplea printed circuit board (PCB). Each MMIC may have a plurality oftransmission channels TX01, TX02, TX03, etc. and a plurality ofreception channels RX01, RX02, RX03, etc. (even if not all channels aredepicted in FIG. 5). For the operation of the radar system, it isimportant that the LO signals used by the MMICs are coherent. Therefore,the LO signal is generated only in one MMIC, the master MMIC 11, and isforwarded to the slave MMICs 12 and 13.

In the example illustrated, each MMIC has output pins P_(TX01),P_(TX02), P_(TX03), etc. which are assigned to TX channels TX01, TX02,TX03, etc. The input pins P_(RX01), P_(RX02), P_(RX03), etc. areassigned to RX channels RX01, RX02, RX03, etc. (P_(RX02) and P_(RX03)are not shown in FIG. 5). The output pins P_(TX01), P_(TX02), P_(TX03),etc. and the input pins P_(RX01), P_(RX02), P_(RX03), etc. can each becoupled to transmission and reception antennas, respectively. If only asingle MMIC is used, all RX and TX channels of the MMIC can be coupledto antennas. If—as in the case illustrated in FIG. 5—a plurality ofMMICs are coupled, one MMIC is operated as the master and the remainingMMICs are operated as the slave. The master MMIC 11 generates the LOsignal for all slave MMICs 12, 13. In the example illustrated, the LOsignal s_(LO)(t) generated in the master MMIC 11 is output at an outputpin, for example the pin PTX03, and is supplied to the slave MMIC 12 viaa line. The slave MMIC 12 receives the LO signal via the line (forexample a stripline arranged on the printed circuit board) at an inputpin. In order to avoid a separate input pin for the LO signal s_(LO)(t),the output pin P_(TX01) of the TX channel TX01 is configured as an inputpin in the present example (and is therefore denoted P_(TX01)′). Themanner in which an output pin of a TX channel can be configured as aninput pin is explained in more detail later. Alternatively, anadditional input pin may be provided for the purpose of receiving anexternally generated LO signal, which is undesirable in someapplications, however.

A slave MMIC can also forward “its” LO signal to a further slave, whichmakes it possible to cascade a master MMIC and a plurality of slaveMMICs. In the example illustrated in FIG. 5, the master MMIC 11generates the LO signal and forwards it to the slave MMIC 12 (via itspin P_(TX0)′) via the output pin P_(TX03). In the same manner, the slaveMMIC 12 forwards the LO signal (received from the master MMIC 11) to afurther slave MMIC 13, as a result of which a plurality of MMICs can beconnected in series (cascaded). The clock signal s_(CLK)(t) is likewiseforwarded from MMIC to MMIC, but separate clock pins are provided forthis purpose (also see FIGS. 9c and 15). The (system) clock signals_(CLK) has a clock frequency of several MHz, whereas the LO signal hasan LO frequency f_(LO) of several GHz (for example 76-81 GHz). The clockgenerator (not illustrated) which generates the clock signal s_(CLK) canbe integrated, for example, in the master MMIC 11 (cf., for example,FIGS. 9c and 15) or can be arranged in a separate chip. In this case,the clock generator may contain an oscillating crystal, for example. Theclock signal s_(CLK) or a clock signal derived therefrom may be, forexample, a reference clock for the local oscillators arranged in theMMICs 11, 12, 13 (cf. FIG. 4, local oscillator 101).

The transmission of the LO signal s_(LO)(t) from one MMIC to the nextMMIC is associated with a delay time which depends, inter alia, on thelength of the lines between the MMICs. In the example illustrated inFIG. 5, the propagation delay of the LO signal s_(LO)(t) between MMIC 11and MMIC 12 is denoted τ₁ and the propagation delay of the LO signals_(LO)(t) between MMIC 12 and MMIC 13 is denoted r2. The propagationdelays τ₁ and τ₂ each correspond to a line length l₁ and l₂ and to aphase shift φ₁ and φ₂, wherein the phase shift φ is generallyproportional to the delay time τ (φ=2π·f_(LO)·τ).

As already mentioned, the phase angle of the LO signals s_(LO)(t)supplied to the slave MMICs is important for carrying out exact radarmeasurements. For example, it would be desirable for the mixers toreceive the LO signal s_(LO)(t) with a defined phase at the referenceinput in the RX channels (see FIG. 4, mixer 104). In some exemplaryembodiments (for example if a plurality of TX channels are used in onechip), it may also be desirable to simultaneously synchronize thetransmission signals supplied to the TX channels.

FIG. 6 illustrates one example of a radar system having a plurality ofMMICs which are arranged on a printed circuit board and are coupled insuch a manner that (in theory) all RX channels in all MMICs (that is tosay the mixers in the RX channels) “see” the LO signal with the samephase angle. According to the example from FIG. 6, the master MMIC 11provides the LO signal s_(LO)(t) at an output pin P_(TX03) which, in thepresent example, is assigned to the output channel TX03 (in a similarmanner to the example from FIG. 5). The output channel TX03 is thereforenot connected to an antenna in order to emit a radar signal, but ratheris used to transmit the LO signal SL(t) to the slave MMIC. Even thoughthe use of an output channel to transmit the LO signal reduces thenumber of available channels, this has the advantage that a designchange is not required since the TX channel is designed for RFfrequencies and a dedicated RF signal output pin therefore does not haveto be provided. In addition, scalability of the radar system can beachieved by using the TX channel to transmit the LO signal. However, ifthe MMIC is not used in a cascaded arrangement (for example in astand-alone arrangement), the TX channel can be used as a complete TXchannel for transmitting radar signals to an antenna. The output pinP_(TX03) of the master MMIC 11 is connected to an input of an RFsplitter 150 which may be arranged on the printed circuit board, likethe MMICs. The line between the pin P_(TX03) and the RF splitter causesa propagation delay of τ_(3,1).

The RF splitter splits the LO signal s_(LO)(t) and provides an LO signalat the splitter outputs for each MMIC. In this case, one of the splitteroutputs is coupled to a feedback input of the master MMIC 11 which isdenoted TX01′ (alternatively FB) and via which the LO signal s_(LO)(t)is fed back to the master MMIC 11. The further splitter outputs areconnected to the corresponding input pins of the slave MMIC, whereinonly the slave MMIC 12 is illustrated in FIG. 6. The propagation delayfrom the RF splitter 150 to the master MMIC 11 is denoted τ_(3,2) andthe propagation delay from the RF splitter 150 to the slave MMIC 12 isdenoted τ_(3,3). If the lines from the outputs of the splitter 150 tothe MMICs (i.e. master MMIC 11, slave MMIC 12, etc.) cause the samepropagation delay (τ_(3,2)=τ_(3,3)), the LO signal arrives at the RXchannels in all MMICs with (in theory) the same phases.

FIG. 7 shows an exemplary implementation of the TX channels of an MMIC(for example the master MMIC 11) which enables coupling to furtherMMICs, as illustrated in FIG. 6. The MMIC 11 therefore comprises a localoscillator 101 (LO) which generates the LO signal s_(LO)(t). This LOsignal is supplied, on the one hand, to an input of a first RFswitch/splitter 110 and, on the other hand, to an input of a second RFswitch/splitter 111. The RF switches/splitters are substantiallysplitter components with selectable inputs which are respectivelydenoted a and b in the figures. Depending on the position of the(electronic) switch, the signal applied to the input a or the signalapplied to the input b is forwarded to the outputs. The control signalsfor the electronic switches are not illustrated for the sake ofsimplicity. In the present example, the RF switch/splitter 111 isconnected in such a manner that input b is selected and the LO signals_(LO)(t) is forwarded to the TX channels TX01, TX02, TX03, etc. Theindividual TX channels can be implemented as illustrated in FIG. 4, forexample.

In FIG. 7, the output of the TX channel TX03 (in a similar manner to theexample from FIG. 6) is not connected to an antenna, but rather to anexternal splitter 150 which can be arranged on the same carrier as theMMIC 11. The LO signal s_(LO)(t) provided at the outputs of the splitter150 is supplied to the various slave MMICs, wherein one output of thesplitter 150 is fed back to a feedback pin P_(FB) of the master MMIIC.The feedback channel FB coupled to the feedback pin P_(FB) in the MMIC11 is designed to pass the fed-back LO signal s_(LO)(t) to a secondinput (input a) of the first RF switch/splitter 110. The feedbackchannel FB may have, for example, a buffer amplifier (LO buffer) whichis not illustrated in the figures for the sake of clarity. One output ofthe first RF switch/splitter 110 can be connected to an input (input a)of the second RF switch/splitter 111, whereas the remaining outputs ofthe first RF switch/splitter 110 provide the LO signals for the RXchannels. The propagation delays τ_(3,1) and τ_(3,2) illustrated in FIG.7 correspond to the situation illustrated in FIG. 6.

As mentioned, FIG. 7 shows the master MMIC 11. The slave MMIC 12 (cf.FIG. 6) can have an identical structure, but the other input (input a)is selected in the second RF switch/splitter 111 in the slave MMIC 12.In this case, the internal LO 101 of the slave MMIC 12 is not used, butrather the LO signal s_(LO)(t) fed back via the feedback channel FB issupplied both to the RX channels (via RF switch/splitter 110) and to theTX channels (via RF switch/splitter 111) of the slave MMIC 12. In thecase of “stand-alone” operation of the MMIC 11 (without a connectedslave MMIC), the input b is selected in the first RF switch/splitter 110and the input b is likewise selected in the second RF switch/splitter111 and the feedback channel FB is inactive. Chips of the same type cantherefore be used in a cascaded radar arrangement in which the RFswitch/splitter 111 transmits the LO signal to another chip. However, italso allows the chips to be used in a stand-alone arrangement in whichall TX channels are used to transmit signals to antennas. In the case ofthe “stand-alone” operation of the MMIC 11, the switch position of thesecond RF switch/splitter 111 is strictly speaking not relevant sincethe LO signal from the local oscillator 101 is forwarded in bothpositions.

FIG. 8 shows a similar example to FIG. 7, but at least one TX channel(for example the TX channel TX01) can be configured as a feedbackchannel. The TX channel TX01 configured as a feedback channel is denotedTX01′ and receives, from the outside, the LO signal which wastransmitted to the RF splitter 150 via the TX03 channel, wherein thesignal output by the RF splitter is passed back to the TX01 channelagain. This configurability of the TX channels enables more flexible useof the MMIC 11 and the additional feedback pin P_(FB) is not required.Apart from the missing separate feedback channel (feedback pin P_(FB)),the MMICs 11, 12 in FIG. 8 are very similar to the MMIC 11 from FIG. 7.In some exemplary embodiments, if feedback of the LO signal is provided,the LO signal generated by the local oscillator 101 cannot be directlypassed to the TX channels, but rather can be first externally fed to theoutside to the splitter 150 and can then be fed back into the chipagain, with the result that a delayed LO signal (in comparison with theLO signal directly present at the local oscillator 101) is ultimatelyused to generate the TX signal output to the antenna.

According to the example from FIG. 8, the MMIC 11 comprises a localoscillator 101 (LO) which generates the LO signal s_(LO)(t). This LOsignal is supplied, on the one hand, to an input of a first RFswitch/splitter 110 and, on the other hand, to an input of a second RFswitch/splitter 111. As in the example according to FIG. 7, the RFswitches/splitters are substantially splitter components with selectableinputs which are respectively denoted a and b. Depending on the positionof the (electronic) switch, the signal present at the input a or thesignal present at the input b is forwarded to the outputs. The controlsignals for the electronic switches are not illustrated for the sake ofsimplicity. In the present example, the second RF switch/splitter 111 inthe master MMIC 11 is connected in such a manner that input b isselected and the LO signal s_(LO)(t) is forwarded from the LO 101 to theTX channels TX01, TX02, TX03, etc. In contrast, in the slave MMIC 12,the second RF switch/splitter 111 is connected in such a manner thatinput a is selected and the LO signal supplied from the outside via thechannel TX01′ is forwarded to the TX channels. Like in the previousexample from FIG. 7, the first RF switch/splitter 110 (in both MMICs 11,12) is connected (switch position a) in such a manner that the LO signalsupplied from the outside via the channel TX01′ is forwarded to the RXchannels and also to an input of the second RF switch/splitter 111. Theswitch position b of the first RF switch/splitter 110 is required onlyfor stand-alone operation in which an external LO signal is not suppliedto the MMIC.

The text below explains how a TX channel can be configured as a feedbackchannel for feeding in an external LO signal. This explanation relatesto both the master MMIC 11 and the slave MMIC 12. Therefore, aconfigurable TX channel (the TX channel TX01′ in the present example)has an output path and an input path. An RF power amplifier 102 isarranged in the output path (in a similar manner to the example fromFIG. 4). A buffer amplifier 105 (LO buffer) is arranged in the inputpath. The output of the RF power amplifier 102 and the input of thebuffer amplifier 105 are connected to the pin P_(TX01). Only one of thesignal paths (input path or output path) is ever active. For thispurpose, the RF power amplifier 102 and the buffer amplifier 105 can bealternately deactivated, with the result that the RF power amplifier 102is active only when the buffer amplifier 105 is inactive and vice versa.The control signals for activating and deactivating the RF poweramplifier 102 and the buffer amplifier 105 are not illustrated for thesake of simplicity.

If an LO signal s_(LO)(t) is supplied to an MMIC from the outside viathe pin P_(TX01)′, the relevant channel (TX01′ in the illustratedexample) must be configured as an input and the buffer amplifier 105 isactive, while the RF power amplifier 102 is inactive. The LO signals_(LO)(t) supplied to P_(TX01)′ is passed, via the buffer amplifier 105,to the input of a splitter, in the present case to the input a of the RFswitch/splitter 110 which forwards the received LO signal to the RXchannels. If a TX channel is not configured as an input, the bufferamplifier 105 is inactive and only the RF power amplifier 102 of therelevant TX channel is active and the TX channel operates in theconventional manner (cf. FIG. 4) and can be connected to an antenna, forexample. The TX channel TX01 can therefore be configured, with theresult that it can be used either as a TX channel which externallyoutputs a signal (either to the antenna or as an LO signal to anotherchip), or receives (if configured as an input) an LO signal from theoutside. The TX channel TX01 can therefore be configured in abidirectional manner.

As can be seen in FIG. 8, the master MMIC 11 and the slave MMIC 12 havea substantially identical structure. Only the switch position of thesecond RF switch/splitter 111 in the slave MMIC 12 differs from that inthe master MMIC 11. Since the LO signal s_(LO)(t) generated by the localoscillator 101 is intended to be passed directly to the TX channels inthe master MMIC 11, the input b is selected in the second RFswitch/splitter 111. The LO signal supplied from the outside (fed backvia the splitter 150) is divided only among the RX channels using thefirst RF switch/splitter 110. In the slave MMIC 12, the LO signalsupplied from the outside (via the splitter 150) is intended to be usedboth for the TX channels and for the RX channels. For this reason, theinput a is selected in the second RF switch/splitter 111 in the slaveMMIC 12.

If the propagation delays τ3,2 and T3,3 are the same, the RX channels ofthe master MMIC 11 and of the slave MMIC 12 “see” the LO signals_(LO)(t) (in theory) with the same phase. Although this equality of thephases applies in theory to a symmetrical connection (that is to sayτ_(3,2)=τ_(3,3)= . . . ) of the master MMIC and slave MMICs according toFIGS. 7 and 8, it is difficult to achieve with sufficient accuracy inpractice since the propagation delays τ_(3,2) and τ_(3,3)—and thereforealso the phase—of the LO signals arriving at the RX channels can changeover the course of time on account of temperature changes. As mentioned,unknown phase changes can have a negative effect on the result of theradar measurements. For this reason, it may be useful to measure, inaddition or as an alternative to the symmetrical connection between themaster MMIC and the slave MMIC according to FIGS. 7 and 8, thepropagation delay (or the phase shift) of the LO signal s_(LO)(t)between two coupled MMICs and to then take into account the measureddelay in the radar measurement. The propagation time determinationdescribed below can therefore be carried out both in a symmetricalstructure and in a non-symmetrical structure, that is to say a structurein which the RF splitter 150 is not used, for example.

FIGS. 9A and 9B show examples of an MMIC having a plurality of TX and RXchannels, wherein the TX channels are designed to determine apropagation delay τ₃ or a phase shift between a first MMIC (for examplemaster MMIC 11) and a second MMIC (for example slave MMIC 12), whereinthis determination of the propagation delay can be carried out in abidirectional manner (from the first MMIC to the second MMIC and viceversa) in order to eliminate asymmetries on account of differenttemperatures of the MMICs and on account of delays of the clock signals_(CLK)(t) between the MMICs. FIG. 9A relates to the case oftransmitting signals from the MMIC 11 to the MMIC 12 via the line L3 andFIG. 9B relates to the reverse case of transmitting signals from theMMIC 12 to the MMIC 11. The MMIC 11 in FIGS. 9A-9C has substantially thesame structure as in the previous example according to FIG. 8, whereinthe TX channels each have two demodulators 115 and 116 and a coupler 117in FIGS. 9A-9C.

In the TX channels, the coupler 117 is connected between the RF poweramplifier 102 and the output pin of the respective TX channel (pinsP_(TX01) and P_(TX03) in FIGS. 9A-9C). The coupler is also connected tothe demodulators 115 and 116 in such a manner that a part (of the power)of the output signal from the relevant TX channel is supplied to the RFinput of the demodulator 115 (that is to say a part of the output signalfrom the amplifier 102) and a part (of the power) of the signal arrivingat the output pin of the respective TX channel is supplied to the RFinput of the demodulator 116. The LO signal s_(LO)(t) is supplied to thereference inputs of the demodulators 115 and 116 via the splitter 110.The master MMIC and the slave MMIC may have a substantially identicalstructure. For the sake of clarity, only the TX channel TX03 isillustrated in more detail for the master MMIC 11 and only the TXchannel TX01 is illustrated in more detail for the slave MMIC 12 inFIGS. 9A and 9B. In the example from FIGS. 9A-9C, the TX channels can beconfigured as an input, like in FIG. 8, but this is optional in thisexample. Therefore, the feedback paths with the buffer amplifiers 105are depicted using dashed lines.

In the present example, the output signal from the RF amplifier 102 ofthe master MMIC 11 is denoted s_(RF,MA)(t) and the output signal fromthe RF amplifier 102 of the slave MMIC 12 is denoted s_(RF,SL)(t).Similarly, the LO signal from the master MMIC 11 is denoted s_(LO,MA)(t)and the LO signal from the slave MMIC 12 is denoted s_(LO,SL)(t). Thesignals s_(RF,MA)(t) and s_(RF,SL)(t) are each amplified andphase-shifted versions of the LO signals s_(LO,MA)(t) and s_(LO,SL)(t).The signal from the master MMIC 11 arriving at the slave MMIC 12 isdenoted s_(RF,MA)′(t) (see FIG. 9A). Similarly, the signal from theslave MMIC 12 arriving at the master MMIC 11 is denoted s_(RF,SL)′(t)(see FIG. 9B). The phase shift between s_(RF,MA)(t) and s_(RF,MA)′(t) isdetermined substantially by the propagation delay τ₃. Similarly, thephase shift between s_(RF,SL)(t) and s_(RF,SL)′(t) is determinedsubstantially by the propagation delay τ₃. In addition to thepropagation delay τ₃, delays of the clock signals s_(CLK) (see FIG. 5,clock s_(CLK) in the MMIC 12 is slightly delayed in comparison with theclock s_(CLK) in the MMIC 11) and temperature differences in theindividual MMICs 11, 12, etc. can also play a role since temperaturechanges even inside the individual MMICs can result in changes in thephases.

The output signals from the demodulators 115 and 116 in the MMICs 11 and12 are supplied to a control unit (for example a microcontroller, notshown in FIG. 9A). The output signals from the demodulators are suppliedto the control unit (possibly by means of analog-to-digital converters),and the control unit is designed to carry out the calculations needed todetermine the propagation delay τ₃. For this purpose, the control unitmay have a processor which can be programmed by means of softwareinstructions to carry out the calculations mentioned. The control unitcan be arranged either in one of the MMICs (for example the master MMIC11) or in a separate control chip (for example a microcontroller whichcan be arranged on the same printed circuit board as the MMICs 11, 12).The MMICs 11, 12 and the control unit can be coupled to a serial dataline, for example, for the purpose of interchanging digital data.

There are various possible ways of determining the propagation delay τ₃and the associated phase shift with the aid of the circuit configurationillustrated in FIGS. 9A-9C. According to the present example, the localoscillator 101 in the master MMIC 11 is used to generate a chirp signalas an LO signal s_(LO,MA)(t) (frequency ramp). In the master MMIC 11,this chirp signal is passed, on the one hand, to the reference input ofthe demodulator 115 via the splitter 110 and is passed (inter alia) tothe RF amplifier 102 in the TX channel TX03 via the splitters 110 and111. The output signal s_(RF,MA)(t) from the amplifier 102 is passed, onthe one hand, to the RF input of the demodulator 115 and, on the otherhand, to the pin P_(TX03) of the master MMIC 11 via the coupler 117. Thesignal output at the pin P_(TX03) of the master MMIC 11 arrives at thepin P_(TX01) of the slave MMIC 12 as a delayed signal s_(RF,MA)′(t) andis forwarded in the slave MMIC 12 to the demodulator 116 via the coupler117. This signal flow corresponds to the case shown in FIG. 9A. In FIG.10, the LO signal (chirp signal) s_(LO,MA)(t) is depicted as a solidline, the signal s_(RF,MA)(t) is depicted as a dash-dotted line and thesignal s_(RF,MA)′(t) is depicted as a (thick) dashed line (the thindashed line is discussed later).

As illustrated in FIG. 10, the internal propagation delay in the masterMMIC 11 is τ_(i) (that is to say the propagation time from the LO 101 tothe splitter 110) and the external propagation delay from the coupler117 in the master MMIC 11 to the coupler 117 in the slave MMIC 12 is τ₃.The entire propagation delay is therefore τ_(i)+τ₃. The differentialfrequency Δf_(M1) is determined with the aid of the demodulator 115 inthe master MMIC 11 and the differential frequency Δf_(S1) (beatfrequency) is determined with the aid of the demodulator 116 in theslave MMIC 12. These differential frequencies Δf_(M1) and Δf_(S1) can beeasily converted into corresponding delay times τ_(i) and τ_(i)+τ₃ bymeans of the ramp gradient K=df/dt (in Hz/s) of the frequency ramp. Inthe present example, the propagation delay τ₃ results from the equation:

τ₃=(Δf _(S1) −Δf _(M1))/K.  (1)

The graph from FIG. 11 illustrates the converse case according to FIGS.9A-9C. According to the present example, the local oscillator 101 in theslave MMIC 12 is used to generate a chirp signal as an LO signals_(LO,SL)(t) (frequency ramp). In the slave MMIC 12, this chirp signalis passed, on the one hand, to the reference input of the demodulator115 via the splitter 110 and is passed (inter alia) to the RF amplifier102 in the TX channel TX01 via the splitters 110 and 111. The outputsignal s_(RF,SL)(t) from the amplifier 102 is passed, on the one hand,to the RF input of the demodulator 115 and, on the other hand, to thepin P_(TX01) of the slave MMIC 12 via the coupler 117. The signal outputat the pin P_(TX01) of the slave MMIC 12 arrives at the pin P_(TX03) ofthe master MMIC 11 as a delayed signal s_(RF,SL)′(t) and is forwarded inthe master MMIC 11 to the demodulator 116 via the coupler 117. Thissignal flow corresponds to the case shown in FIG. 9B. In FIG. 11, the LOsignal (chirp signal) s_(LO,SL)(t) is depicted as a solid line, thesignal s_(RF,SL)(t) is depicted as a dash-dotted line and the signals_(RF,SL)′(t) is depicted as a (thick) dashed line.

As illustrated in FIG. 11, the internal propagation delay in the slaveMMIC 12 is τ_(i) (that is to say the propagation time from the LO 101 tothe splitter 110) and the external propagation delay from the coupler117 in the slave MMIC 12 to the coupler 117 in the master MMIC 11 is τ₃(the entire propagation delay is τ_(i)+τ₃). The differential frequencyΔf_(S2) is determined from the demodulated baseband or intermediatefrequency signal with the aid of the demodulator 115 in the slave MMIC12 and the differential frequency Δf_(M2) (beat frequency) is determinedfrom the demodulated baseband or intermediate frequency signal with theaid of the demodulator 116 in the master MMIC 11. As in the previouscase, these differential frequencies Δf_(S2) and Δf_(M2) can beaccordingly easily converted into corresponding delay times τ_(i) andτ_(i)+τ₃ by means of the ramp gradient K of the frequency ramp. In thepresent example, the propagation delay τ₃ results from the equation(similar to equation 1):

τ₃=(Δf _(M2) −Δf _(S2))/K.  (2)

In theory, the same value would have to be calculated in both cases (forboth signal flow directions) for the propagation delay τ₃. However, inpractice, it is not possible to disregard the fact that the frequencyramps (LO signals) s_(LO,MA)(t) and s_(LO,SL)(t) cannot be triggered atthe same time, but rather the frequency ramp in the slave MMIC 12 alwayslags behind the corresponding frequency ramp in the master MMIC 11 onaccount of a propagation delay τ_(CLK) of the clock signal s_(CLK) (seeFIG. 5), also referred to as a clock delay. As illustrated in FIG. 6,the clock signal is distributed to the individual MMICs via a separateline. The clock signal therefore has a propagation delay which differsfrom the LO signal in the slave MMIC. In FIG. 10, the thin dashed lineillustrates the signal s_(LO,SL)(t) in the slave MMIC 12 and, in FIG.11, the thin dashed line illustrates the signal s_(LO,MA)(t) in themaster MMIC 11. The clock delay results in the differential frequencyΔ_(M2) (see FIG. 11) in the master MMIC 12 being systematically measuredto be too large and the differential frequency Δf_(S1) (see FIG. 10) inthe slave MMIC 12 being systematically measured to be too small. As aresult, the effect of the clock delay is very similar to the effect ofthe Doppler shift when measuring the distance using an FMCW radarsystem. The systematic error mentioned can be eliminated by forming theaverage of the propagation time measurement in both directions(corresponding to FIGS. 9A and 9B). Consequently, the propagation delayτ₃ results from averaging equations 1 and 2:

τ₃=(Δf _(S1) +Δf _(M2) −Δf _(M1) −Δf _(S2))/(2K).  (3)

The clock delay τ_(CLK) can be determined in a similar manner from thedifference between equations 1 and 2. The phase shift φ₃ belonging tothe propagation delay τ₃ follows from the equation φ₉₃=2π·f_(LO)·τ₃.

An alternative approach to determining the propagation delay τ₃ and thephase shift φ₃ with the aid of the system from FIGS. 9A-9C is explainedbelow. In contrast to the previously described example in which chirpsignals are used as LO signals for measuring the phase shift φ₃ (and thepropagation delay τ₃), LO signals at a defined frequency are used in thefollowing example. The following example can also be carried out using asystem according to FIGS. 9A-9C and differs from the previouslydescribed example only in the evaluation of the signals.

In the present example, the local oscillator in the master MMIC 11 isoperated at a frequency f_(LO,MA)=f_(LO)+Δf, whereas the localoscillator in the slave MMIC 12 is operated at a frequencyf_(LO,MA)=f_(LO). The differential frequency or the frequency offset Δfshould be selected to be so small that the associated wavelength λ isgreater than the line length L3 between the MMICs 11 and 12. Thewavelength is calculated according to λ=c/Δf, where c is the propagationspeed of the LO signals via the line arranged on the printed circuitboard between the MMICs 11 and 12. Since the propagation speed and theapproximate line length L3 are known, the frequency offset can be setaccordingly. Therefore, Δf<c/L3.

First of all, Δf=Δf₁ is selected as the frequency offset. If the signalflow direction from the master MMIC 11 to the slave MMIC 12 isconsidered (see FIG. 9A), the phase of the outgoing signal s_(RF,MA)(t)at the output of the TX channel TX03 (that is to say at the coupler 117in the master MMIC 11) is measured by means of the demodulator 115, fromwhich the propagation delay Ti inside the chip can again be determinedin a similar manner to that in the previous example. The signals_(RF,MA)′(t) transmitted by the master MMIC 11 and arriving at theslave MMIC 12 is supplied to the demodulator 116 in the slave MMIC 12via the coupler 117 in the TX channel TX01, wherein the demodulation inthe slave MMIC 12 is carried out using the LO signal s_(LO),SL(t)(frequency f_(LO,SL)=f_(LO,MA)−Δf₁). The demodulator 116 provides, atits outputs, an output signal having the differential frequency Δf₁ andthe associated phase

ψ₁=2π·Δf ₁·(τ₁+τ₃),  (4)

from which the propagation delay τ₃ can be easily determined. Thissituation is illustrated in the upper graph in FIG. 12 in which thesignal generated in the master MMIC 11 and demodulated in the slave MMIC12 is illustrated. In order to verify this measurement, it is repeatedwith a different frequency offset Δf=Δf₂ (cf. lower graph in FIG. 12).In the present example, the frequency offset is reduced by 25%, as aresult of which the phase

ψ₂=2π·Δf ₂·(τ_(i)+τ₃)  (5)

is also accordingly reduced at the output of the modulator 112 in theslave MMIC 12 (see FIG. 9A) since the propagation delay (τ_(i)+τ₃)remains the same. If the phase ψ₂ does not change in proportion to thefrequency, a sudden phase change from 360 to zero degrees has occurredand the measurement is not unambiguous. This can happen when thefrequency offset Δf does not match the line length L3.

As in the previous example, the measurement can also be carried out in abidirectional manner in the present case and the signal flow from theslave MMIC 12 to the master MMIC 11 can also be considered (see FIG.9B). Errors which are caused by dissimilar on-chip propagation delays inthe MMICs 11 and 12, for example, can be compensated for by averagingthe measurement results obtained for different signal flow directions.

More reliable results are obtained using a differential approach. Thatis to say, only phase changes are considered. If the difference betweenequation 5 and equation 4 is formed, the following is obtained

ψ₂−ψ₁=2π·(Δf ₂ −Δf ₁)·(τ_(i)+τ₃).  (6)

In the case of a multiplicity of measurements with different frequencyoffsets, a multiplicity of equations similar to equation 6 are obtained,which equations in practice usually form an overdetermined system ofequations since the equations will not be linearly dependent on accountof noise, measurement errors, etc. This system of equations

ψ_(k)−ψ₁=2π·(Δf _(k) −Δf ₁)·(τ_(i)+τ₃), for k=2, . . . N,  (7)

can be solved using known methods in order to obtain a solution for thepropagation delay τ_(i)+τ₃, for example on the basis of the leastsquares concept. In another example, the results for the propagationdelay τ_(i)+τ₃ can be averaged.

FIG. 9C is a simplified version of the example from FIG. 9B in which allof those components which are not required for the measurement accordingto equations 6 and 7 have been omitted. As mentioned, the localoscillators in the two MMICs 11 and 12 are set to different frequencies,wherein the local oscillator in the slave MMIC 12 oscillates at afrequency F_(LO,SL) which is lower than the frequency f_(LO,MA) of thelocal oscillator in the master MMIC 11 by a frequency offset Δf_(k). Theamplified (and phase-shifted owing to the propagation delay τ_(i) insidethe chip) local oscillator signal s_(LO,SL)(t) is output by the slaveMMIC 12 as an RF signal s_(RF,SL)(t) at the chip contact P_(TX01) (pinor solder ball) and is transmitted to the chip contact P_(TX03) of themaster MMIC 11 via the line L3. The RF signal s_(RF,SL)′(t) arriving atthe master MMIC 11 is additionally phase-shifted on account of thepropagation delay τ₃ of the line L3 and is supplied, via the coupler117, to the demodulator 116 which demodulates (demodulator 116) thesignal s_(RF,SL)′(t) using the local oscillator signal s_(LO,MA)(t)generated in the master MMIC 11. As mentioned, the demodulator 116provides measured values for the frequency offset Δf_(k) and, inparticular, for the phase ψ_(k). These measured values can be evaluatedaccording to equation 7 in order to calculate the propagation delayτ_(i)+τ₃. The propagation delay Ti inside the chip can be separatelymeasured, as described, and can be considered to be known for thecalculation of τ₃. The measurement can be repeated for a multiplicity offrequency offsets Δf_(k), in which case the master MMIC 11 and the slaveMMIC 12 swap roles.

FIG. 13 illustrates a master MMIC 11 which is substantially identical tothe example from FIG. 9A but has an additional phase modulator 106connected upstream of the RF power amplifier 102 in the TX channels. Thefrequency offset Δf mentioned above with respect to FIG. 12 can also beachieved by means of phase modulation of the LO signal s_(LO,MA)(t)instead of changing the frequency of the local oscillator itself. Thismodulation frequency Δf appears at the output of the demodulator duringdemodulation in the slave MMIC 12 and the associated phase shiftprovides information on the propagation delay τ₃ sought.

FIG. 14 uses a flowchart to illustrate a general example of a method fordetermining the propagation delay or phase shift of an LO signal duringtransmission from a first (for example master) MMIC to a second (forexample slave) MMIC or vice versa. This method can be carried out, forexample, using the radar systems which are illustrated in FIGS. 7-9 and13 and have a plurality of coupled MMICs. According to FIG. 14, themethod comprises generating a first RF oscillator signal in a first chip(see, for example, FIG. 9A, LO signal s_(LO,MA)(t) in MMIC 11). Thisfirst RF oscillator signal is supplied to a TX channel of the first chip(see FIG. 14, step S1). At the output of the TX channel, the first RFoscillator signal already has a phase shift on account of thepropagation delay inside the chip. In FIG. 9A, this phase-shifted firstRF oscillator signal at the output of the TX channel TX03 is denoteds_(RF,MA)(t). The method also comprises generating a second RFoscillator signal in a second chip (see FIG. 14, step S3, and, forexample, FIG. 9A, LO signal s_(LO,SL)(t) in MMIC 12). The first RFoscillator signal is transmitted from the first chip to the second chipvia a transmission line (see FIG. 14, step S3). This transmissionresults in a propagation delay and therefore also a phase shift. In FIG.9A, this first RF oscillator signal arriving at the MMIC 12 is denoteds_(RF,MA)′(t). The method finally comprises determining a propagationdelay of the first RF oscillator signal arriving at the second chip bymeans of demodulation using the second RF oscillator signal (see FIG.14, step S3). In FIG. 9A, this demodulation is carried out by thedemodulator 116 in the slave MMIC 12, for example. In the examplesdescribed above, the propagation delay sought is denoted τ₃ and can bedetermined according to equation 3 or from the phase shift illustratedin FIG. 12, for example.

As already mentioned with respect to FIGS. 9A and 9B, the method can becarried out in a bidirectional manner. In this case, the signal from thesecond RF oscillator is supplied to a TX channel of the respective otherchip, with the result that the second RF oscillator signal from thesecond chip (cf. FIG. 9B, MMIC 12) is transmitted to the first chip (cf.FIG. 9B, MMIC 11) via the same transmission line as in the reverse case.A propagation delay of the second RF oscillator signal arriving at thefirst chip is determined by means of demodulation using the first RFoscillator signal. In FIG. 9B, this demodulation is carried out by thedemodulator 116 in the master MMIC 11, for example. The actualpropagation delay can be determined on the basis of the two propagationdelay values previously determined for different signal flow directions,for example by means of averaging according to equation 3.

The first and second RF oscillator signals may be either chirp signals(as illustrated in FIGS. 10 and 11, for example) or RF signals with anadjustably constant frequency (as illustrated in FIG. 12, for example).In the case of chirp signals, the first RF oscillator signal (forexample generated in the master MMIC) contains at least one frequencyramp and the second RF oscillator signal (for example generated in themaster MMIC) contains at least one corresponding frequency ramp. Asmentioned above, two corresponding frequency ramps may have a delayrelative to one another, which delay corresponds to the clock delay. Inthis case, the first propagation delay is determined on the basis of afirst beat frequency which results from the demodulation of the first RFoscillator signal arriving at the second chip using the second RFoscillator signal. The propagation delay is effected for the reversesignal flow direction and the effect of the clock delay can beeliminated by means of averaging. Furthermore, the average clock delaycan be determined by averaging the difference between the first andsecond cases. This procedure is similar to the determination of speedand distance in the FMCW triangulation method. In this method, thedistance would represent the propagation delay τ₃ here and the speedwould represent the clock delay.

When using RF signals with a static (adjustably constant) frequency, thefirst RF oscillator signal may have a first frequency and the second RFoscillator signal may have a second frequency which differs from thefirst frequency by a defined frequency offset. This frequency offset canbe generated by detuning the local oscillator in one of the two MMICs(cf. FIGS. 9A and 9B, oscillator 101) or by means of phase modulation ofone of the two RF oscillator signals (cf example from FIG. 13, phasemodulator 106). In this case, the first propagation delay can bedetermined on the basis of a phase which is assigned to the frequencyoffset and results from the demodulation of the first RF oscillatorsignal arriving at the second chip using the second RF oscillatorsignal. This measurement can be repeated for at least one furtherfrequency offset. For this purpose, the frequency offset can be changedand an accordingly changed phase can be measured (cf. FIG. 12, phases ψ₁and ψ₂). The measurements can again be carried out in a bidirectionalmanner.

Some aspects of the radar systems described here are summarized below.It goes without saying that this is not a complete, but rather only anexemplary, summary of technical features. One example of a radar systemis suitable for implementing the method described above. According tothis exemplary embodiment, the system comprises a first chip having afirst RF contact (cf., for example, FIG. 9A, MMIC 11, pin P_(TX03)) anda second chip having a second RF contact (cf., for example, FIG. 9A,MMIC 12, pin P_(TX0l)). A first RF oscillator is integrated in the firstchip and has an output which is coupled to the first RF contact via atleast one TX channel. A second RF oscillator is integrated in the secondchip. The system also comprises a transmission line (cf., for example,FIG. 9A, line L3 with propagation delay τ₃) which connects the first RFcontact on the first chip to the second RF contact on the second chip.At least one first demodulator is arranged in the second chip. Thisdemodulator has an RF input which is coupled to the second RF contact aswell as a reference input which is coupled to an output of the second RFoscillator. The first RF oscillator is designed to generate a first RFoscillator signal which is transmitted to the RF input of the firstdemodulator via the first RF contact, the transmission line and thesecond RF contact. A control unit (controller) is designed to determinea first propagation delay of the first RF oscillator signal arriving atthe second chip on the basis of information obtained from the firstdemodulator (for example according to equation 3 or on the basis of thephase shifts illustrated in FIG. 12).

Another exemplary embodiment relates to the case of a symmetricallystructured radar system with feedback of the LO signal to the masterMMIC, as illustrated in FIGS. 6 to 8. According to the examplesdescribed here, the radar system comprises a carrier (for example aprinted circuit board, PCB), a first chip and at least one second chipwhich are arranged on the carrier. The first chip has an RF oscillatorwhich is designed to generate an RF oscillator signal and to output itat a first RF output contact. The system also has an RF splitterarranged on the carrier. The RF splitter has an input and a first outputand at least one second output. A first transmission line connects theRF output contact of the first chip to the input of the RF splitter. Asecond transmission line connects the first output of the RF splitter toan RF input of the first chip (cf., for example, FIG. 8, feedback of theRF oscillator signal to the master MMIC 11). A third transmission lineconnects the second output of the RF splitter to an RF input of thesecond chip (cf., for example, FIG. 8, transmission of the RF oscillatorsignal to the slave MMIC 12). In this case, the second and thirdtransmission lines are configured in such a manner that they cause thesame propagation delay during operation when transmitting the RFoscillator signal.

During operation, the RF oscillator signal can be transmitted to the RFsplitter via the RF output contact of the first chip and the firsttransmission line and can be fed back to the first chip via the secondtransmission line and the RF input of the first chip. The first chip maycontain an integrated RF splitter which is designed to forward thefed-back RF oscillator signal to the reception channels contained in thefirst chip. Similarly, the second chip may have an integrated RFsplitter which is designed to forward the RF oscillator signal receivedvia the RF input of the second chip to the reception and transmissionchannels contained in the second chip.

With respect to the examples of radar systems having a plurality ofMMICs, as explained above, it may also be desirable to measure the powerof the LO signal arriving in a slave MMIC (and generated in the masterMMIC). In the example from FIG. 9A and also in the example shown belowin FIG. 15, this LO signal arriving in the slave MMIC 12 is denoteds_(RF,MA)′(t). According to FIG. 15, this arriving LO signals_(RF,MA)′(t) in the slave MMIC 12 is supplied, via the coupler 17 whichis arranged in the vicinity of the chip contact P_(TX01) of the slaveMMIC 12, (inter alia) to a power detector 118 which is designed togenerate and output a measured value representing the power of thesupplied signal (for example as a DC voltage signal). Such RF powerdetectors generally comprise one or more diodes and are known per se andare therefore not explained any further here.

In practice, situations may arise in which the signal power of thesignal s_(RF,MA)′(t) arriving at the slave MMIC 12 is too low for the RFpower detector 118 to be able to generate a reliable measured value(with a sufficient signal-to-noise ratio). In order to improve thesituation, the local oscillator 101 is also activated in the slave MMIC12 and generates the LO signal s_(LO,SL)(t) which is likewise suppliedto the coupler 117, wherein the frequencies f_(LO) of the two LO signalss_(LO,SL)(t) and s_(RF,MA)′(t) are the same for this measurement. Thecoupler 117 is designed in such a manner that a part of the power of theLO signal s_(LO,SL)(t) is likewise forwarded to the RF power detector118, which results in the LO signals s_(LO,SL)(t) and s_(RF,MA)′(t)being superimposed at the input of the RF power detector 118. Theaverage power of the sum signal s_(LO,SL)(t)+s_(RF,MA)′(t) depends onthe phase difference Δϕ of the two signals which can be varied, forexample, using the phase shifter 106 in the channel TX03 of the masterMMIC 11. Additionally or alternatively, a phase shifter in the channelTX01 of the slave MMIC 12 can also be used to vary the phase differenceΔϕ.

It is assumed for the following considerations that the LO signalss_(RF,MA)′(t) and s_(LO,SL)(t) can be modeled as follows:

s _(RF,MA)′(t)=A _(MA) cos(2π·f _(LO) ·t+Δϕ+ϕ _(MA)′),  (8)

s _(LO,SL)(t)=A _(SL) coS(2π·f _(LO) ·t+ϕ _(SL)).  (9)

In this case, the phases ϕ_(MA)′ and ϕ_(SL) can be assumed to be zerowithout restricting generality. The instantaneous power of the sumsignal is proportional to the square of the sum signals_(LO,SL)(t)+s_(RF,MA)′(t), wherein the RF power detector 118 measuresthe temporal average of the instantaneous power. This temporal averageis therefore

P _(LO)=1/T∫ ₀ ^(T)(s _(LO,SL)(t)+s _(RF,MA)′(t))² dt=

=½(A _(MA) ² +A _(SL) ²)+A _(MA) ·A _(SL)·cos(Δϕ),  (10)

whereas the sought power of the LO signal s_(RF,MA)′(t) transmitted fromthe master MMIC 11 to the slave MMIC 12 is equal to A_(MA) ². It isnoted at this point that the power PLO of the sum signal issignificantly higher than the power A_(MA) ² of the signal s_(RF,MA)′(t)alone. Furthermore, the power P_(LO) is a function of the phase shiftΔϕ. A technique of how the power A_(MA) ² can be determined withsufficient accuracy from a number of measured values P_(LO) is presentedbelow, wherein the measured values P_(LO) are recorded in the case ofdifferent phase shifts. In a general example, N measured values P_(LO)are recorded, wherein the phase shift Δϕ is increased by 2kπ/N beforeeach measured value (k is an integer value of greater than or equal to1). In the following example, k=1 (one period) and N=8 are assumed. Inthis case, eight measured values which are uniformly distributed overone period of a cosine curve are obtained. These eight measured valuesare illustrated in FIG. 16, by way of example. It goes without sayingthat the analog output voltage of the power detector 118 can bedigitized before the digital further processing. However, a suitableanalog-to-digital converter is not illustrated in the figures for thesake of clarity.

With k=2 and N=8, eight measured values each with a phase increment of90° (2·2·π/8 rad) and therefore two periods of the cosine curve areobtained. The amplitude A_(MA)·A_(SL) of the cosine termA_(MA)·A_(SL)·cos(Ac) from equation 10 can be calculated from themeasured values in a simple manner using a discrete Fourier transform.If the number of measured values is a power of two (for example 4, 8,16, 32, etc.), the calculation can be carried out efficiently using theFFT algorithm (Fast Fourier Transformation). If the measured valuesrepresent only one period of a cosine curve (k=1), the value sought isthe second value of the spectrum calculated (for example by means ofFFT) (the first value would be the DC component). In the case of twoperiods (k=2), the sought value is the third value of the calculatedspectrum, etc. It is pointed out at this point that the FFT is only oneexample of many possible algorithms. However, the FFT algorithm can beimplemented in a comparatively simple manner in hardware.

FIG. 17 is a flowchart for illustrating one example of a method formeasuring the power of a first RF oscillator signal received in a radarMMIC (see FIG. 17, step S5). The first RF oscillator signal may be, forexample, the LO signal generated in a master MMIC and transmitted to aslave MMIC; the slave MMIC receives the LO signal which is denoteds_(RF,MA)(t) in the example from FIG. 15. In the radar MMIC, a second RFoscillator signal can be generated using a local oscillator, wherein thefirst RF oscillator signal and the second RF oscillator signal have thesame frequency but have an adjustable phase shift relative to oneanother (see FIG. 17, step S6). In the example from FIG. 15, this secondRF oscillator signal is the LO signal s_(LO,SL)(t) generated in theslave MMIC 12. The first RF oscillator signal and the second RFoscillator signal are superimposed at an input of an RF power detector,thus forming the sum signal s_(LO,SL)(t)+s_(RF,MA)(t) (See FIG. 17, stepS7). A multiplicity of measured values representing the power of the sumsignal are generated using the RF power detector (cf. FIG. 15, powerdetector 118), wherein a particular phase shift is assigned to each ofthe measured values (see FIG. 17, step S8). The power of the first RFoscillator signal is then determined on the basis of the multiplicity ofmeasured values (see FIG. 17, step S9). This may comprise the Fouriertransform of the measured values, for example. As already explainedfurther above with reference to FIG. 16, the phase shifts assigned tothe measured values may be equidistant and may be uniformly distributedover one or more full phase rotations (that is to say 3600 correspondsto 27 rad), which allows the further processing of the measured valuesby means of a Fourier transform. If the phase shifts are distributedover one full phase rotation (for example 00°, 45°, 90°, 135°, 180°,225°, and 315°), as in the graph from FIG. 16, the second “bin”(spectral value) of the discrete Fourier transform represents the powervalue sought (cf. equation 10). The first bin represents the DCcomponent, as already mentioned further above.

Although exemplary embodiments have been described and illustrated withreference to one or more implementations, changes and/or modificationscan be made to the examples illustrated without departing from thespirit and scope of the accompanying claims. In particular, with regardto the different functions performed by the components or structuresdescribed above (units, assemblies, apparatuses, circuits, systems,etc.), the designations (including the reference to a “means”) which areused to describe such a component should also correspond to any othercomponent or structure which performs the specified function of thedescribed component (that is to say which is functionally equivalent)even if it is not structurally equivalent to the disclosed structurewhich performs the function in the exemplary implementations describedhere.

What is claimed is:
 1. A radar system, comprising: a first chip, whereinthe first chip has radio frequency (RF) oscillator that is configured togenerate an 1U oscillator signal and to output the RI oscillator signalat a first RE output contact; at least one second chip; an RF splitterhaving an input, a first output, and a second output; a firsttransmission line that connects the first 1′ output contact of the firstchip to the input of the RF splitter; a second transmission line thatconnects the first output of the splitter to an RE input of the firstchip; and a third transmission line that connects the second output ofthe RE splitter to an RE input of the second chip; wherein the secondtransmission line and third transmission line are configured to cause asame propagation delay during operation when transmitting the RFoscillator signal.
 2. The radar system as claimed in claim 1, wherein,during operation, the RF oscillator signal is transmitted to the REsplitter via the first RE output contact of the first chip and the firsttransmission line and is fed back to the first chip via the secondtransmission line and the RF input of the first chip.
 3. The radarsystem as claimed in claim 2, wherein the first chip contains anintegrated RE splitter that is configured to forward the RF oscillatorsignal which has been fed back to reception channels contained in thefirst chip.
 4. The radar system as claimed in claim 2, wherein thesecond chip contains an integrated RF splitter that is configured toforward the RF oscillator signal received via the RF input of the secondchip to reception channels or transmission channels contained in thesecond chip.
 5. A radar system, comprising: a first chip having a firstradio frequency (RF) contact; a second chip having a second RF contact;a first RF oscillator that is integrated in the first chip and has anoutput that is coupled to the first RF contact via at least onetransmission (TX) channel; a second RF oscillator integrated in thesecond chip; a transmission line that connects the first RF contact onthe first chip to the second RF contact on the second chip; at least onefirst demodulator that is arranged in the second chip and has a firstRI; input and a first reference input, wherein the first RF input iscoupled to the second RF contact and the first reference input iscoupled to an output of the second RI; oscillator; wherein the first RFoscillator is configured to generate a first RF oscillator signal thatis transmitted to the first RF input of the at least one firstdemodulator via the first RF contact, the transmission line, and thesecond RF contact, and wherein a controller is configured to determine afirst propagation delay of the first RF oscillator signal arriving atthe second chip on a basis of information obtained from the at least onefirst demodulator.
 6. The radar system as claimed in claim 5, wherein:the second RF oscillator has an output that is coupled to the second RIPcontact at least one further transmission (TX) channel, at least onesecond demodulator is arranged in the first chip and has a second RI;input and a second reference input, wherein the second RF input iscoupled to the first RF contact and the second reference input which iscoupled to an output of the first oscillator, the second RF oscillatoris designed to generate a second RIP oscillator signal that istransmitted to the second RF input of the at least one seconddemodulator via the second RF contact, the transmission line, and thefirst RF contact, and the controller is configured to determine a secondpropagation delay of the second RF oscillator signal arriving at thefirst chip on a basis of information obtained from the at least onesecond demodulator.
 7. The radar system as claimed in claim 6, whereinthe controller is configured to determine a propagation delay assignedto the transmission line on a basis of the first propagation delay andthe second propagation delay.
 8. The radar system as claimed in claim 5,wherein the first RF oscillator signal has at least one frequency rampand the second RF oscillator signal has a corresponding frequency ramp.9. The radar system as claimed in claim 8, wherein the first propagationdelay is determined on a basis of a first beat frequency that resultsfrom a demodulation of the first RF oscillator signal arriving at thesecond chip using the second RF oscillator signal.
 10. The radar systemas claimed in claim 5, wherein the first RF oscillator signal has afirst frequency and the second RF oscillator signal has a secondfrequency that differs from the first frequency by a defined frequencyoffset.
 11. The radar system as claimed in claim 10, wherein the firstpropagation delay is determined on a basis of a phase which is assignedto the frequency offset and results from the demodulation of the firstRF oscillator signal arriving at the second chip using the second RFoscillator signal.
 12. The radar system as claimed in claim 10, wherein,in order to determine the first propagation delay, the controller isconfigured to: determine two phases, which are a result of ademodulation of the first RF oscillator signal arriving at the at leastone first demodulator, for two corresponding frequency offsets,calculate a phase difference from the two phases and an associatedfrequency offset difference, and calculate the first propagation delayon a basis of the phase difference and the associated frequency offsetdifference.
 13. A radar monolithic microwave integrated circuit (MMIC),comprising: a local oscillator (LO) for generating a radio frequency(RF) oscillator signal; and at least one first transmission (TX) channelthat is connected to a pin of the MMIC; wherein the TX channel isconfigured both to output the RF oscillator signal via the pin of thefirst TX channel and to receive a further RF oscillator signal via thepin of the first TX channel.
 14. A method, comprising: receiving a firstradio frequency (RE) oscillator signal via an RF chip contact of a radarchip; generating a second RF oscillator signal by an RE oscillator inthe radar chip, wherein the first RE oscillator signal and the second REoscillator signal have a same frequency and an adjustable phase shiftrelative to one another; generating a sum signal by superimposing thefirst RE oscillator signal and the second RF oscillator signal;generating a plurality of measured values, which represent a power ofthe sum signal, for a plurality of phase shifts, wherein a differentphase shift of the plurality of phase shifts is assigned to each of theplurality of measured values; and determining a value representing apower of the first RF oscillator signal on a basis of the plurality ofmeasured values.
 15. The method as claimed in claim 14, wherein theplurality of phase shifts assigned to the plurality of measured valuesare uniformly distributed over one or more periods of full phaserotations.
 16. The method as claimed in claim 14, wherein determiningthe value representing the power of the first RF oscillator signalcomprises calculating a Fourier transform of the plurality of measuredvalues.
 17. A radio frequency (RF) circuit, comprising: an RE chipcontact of a radar chip configured to receive an external first RFoscillator signal; a local oscillator that is arranged in the radar chipand is configured to generate a second RE oscillator signal, wherein theexternal first RE oscillator signal and the second RF oscillator signalhave a same frequency; a phase shifter configured to set a phase shiftbetween the external first second oscillator and the second oscillatorsignal; and an RE power detector that is arranged in the radar chip andhas an input that is coupled to the RE chip contact and to an output ofthe local oscillator by a coupler; with a result that the first REoscillator signal and the second RE oscillator signal are superimposedat the input of the RE power detector, with a result that the RF powerdetector measures a power of the superimposed signal.
 18. The RF circuitas claimed in claim 17, further comprising: a controller configured tocapture a plurality of measured values, which are provided by the RFpower detector and represent the power of the superimposed signal, for aplurality of phase shifts, wherein a different phase shift of theplurality of phase shifts is assigned to each of the plurality ofmeasured values.
 19. The RF circuit as claimed in claim 17, wherein: theplurality of phase shifts assigned to the plurality of measured valuesare uniformly distributed over one or more periods of full phaserotations, and the controller is configured to calculate a Fouriertransform of the plurality of measured values and to determine a powerof the first RE oscillator signal on a basis of the calculated Fouriertransform.